The present invention relates to radio frequency (RF) power control in a transmitter, particularly a transmitter of a cellular telephone handset.
Many radio transmitter systems require accurate control of the transmitted RF power. This is commonly achieved by a feedback control system in which the difference between the sampled output power from the power amplifier (PA) and a reference control derives an error signal that controls the PA gain. This is a common technique because it is usually far easier to implement accurate power measurement with supply voltage and temperature variation than it is to maintain PA output power within acceptable limits. Additionally, in many radio transmitter applications the output power needs to be accurately set to one of a number of prescribed power levels within some total dynamic range.
TDMA systems usually require the transmitted signal burst to adhere to a strict power versus time template. Thermal heating effects in the PA generally cause PA parameters such as gain and/or efficiency to decrease throughout the burst. A feedback control system can automatically level the transmitted burst, provided of course that there is sufficient control loop bandwidth and power output capability. Also, many TDMA systems require the handset's transmitted power to be smoothly ramped to and from the average burst power. This is commonly specified in order to reduce the amount of adjacent channel interference caused by the power ramp pulse, and/or to limit the power transmitted in adjacent timeslots.
A typical power control loop used in (but not restricted to) mobile radio communications is shown in FIG. 1 of the accompanying drawings. In such a control loop, the transmit power Ptrans output by a power amplifier 11 (which is fed from a voltage controlled oscillator 10 to an antenna 16) is sampled through a coupler 12 and the value detected in a detector 13 (shown here as a Schottky diode) which outputs a related detector voltage Vdet which is then compared in a comparator 14 with a controllable reference R. The difference signal is integrated in an integrator 15 and the integrated difference signal ΔS is fed back and used to adjust the gain of the power amplifier 11 in the appropriate direction in order that the detector voltage Vtrans moves towards the reference voltage R. This is a classic feedback technique and in this manner, the controllable reference voltage sets the power output and can be dynamically controlled to effect the ramp signals required for example in TDMA system handsets. However, the voltage Vdet of a diode detector has a square-law relationship to the transmitted (input) power Ptrans and furthermore at low power levels the exponential characteristic of the Schottky diode dominates the response.
In relation to PA output power variance with respect to control reference, the transfer function of the circuit shown in FIG. 1 is defined solely by the detector's characteristic. Implementations of power detectors are commonly based around either Schottky diodes (as shown) or log-amps. These solutions provide non-linear transfer functions and also have limited dynamic range, especially in the case of Schottky diode detectors. This results in the PA power setting with respect to its control input being non-linear and beyond the control of the equipment manufacturer. The transfer function derivative (gradient) therefore varies with power setting and in implementations using Schottky diodes, the detector usually dominates the overall gradient variation and implicitly the loop bandwidth variation.
Furthermore, variation of control loop bandwidth with power setting is undesirable and results in a varying dynamic response. In GSM (Global Systeme Mobile) systems, which are one form of TDMA system, the ramp profile should ideally be constant with each power setting. Non-linear transfer functions result in the provision of a unique ramp shape at each power setting and this demands extra memory. Moreover, the system is intolerant to unit-to-unit detector and PA variation, as well as power variation (for example over temperature or due to calibration errors).
There is a need therefore to provide a linear control of the RF output power from a handset PA and this is preferably achieved by means of a feedback control system.
According to the present invention therefore there is provided RF transmitter power control system having an oscillator:                a power amplifier receiving the output of the voltage controlled amplifier as a first input, having a control input, and generating an output signal for passing to an antenna;        a sampling device for deriving a signal representative of the output signal;        a multiplier for multiplying the representative signal by itself to provide an output component which is linearly related to the representative signal;        a variable gain amplifier receiving the linear output component of the multiplier as a first input and having a variable control input as a control reference and an output;        a comparator for comparing the output of the variable gain amplifier with a reference and producing a difference signal; and        an integrator for integrating the difference signal and providing it as a control input to the power amplifier.        
The way forward to providing a consistent and predictable overall loop transfer function e.g. loop bandwidth, phase margin, is by definition of linear transfer functions for the individual circuit blocks within the loop. The control loop bandwidth is governed by both the derivatives (rate of change) and absolute gain of the block transfer functions. A linear transfer function gives a constant derivative.
Several advantages result from the use of the system of the invention, particularly for TDMA mobile communications applications:                The use of an analogue multiplier to convert the sampled RF signal to a DC voltage results with a linear relationship between the transmitted RF voltage and the detected voltage. This avoids the linearity problems encountered with the use of Schottky diode detectors or log-amplifiers. The dynamic range of the multiplier detector is only limited by its large signal handling and DC offsets or thermal noise at low input signal powers. The method is identical to a ‘direct conversion receiver’ architecture and implies the dynamic range is significantly greater than existing detector solutions. Furthermore, it is possible to make wideband multiplier circuits that remove any necessity to incorporate switching for operation in different frequency bands, or compensate for the slope in frequency response.        The use of a fixed error detection reference combined with a VGA results in an identical error slope (dV(detected)/dV(transmitted)) for all in-lock RF output power settings. This results in a constant control loop bandwidth provided the PA control characteristic is linear. In practice, this is never the case, but is not the dominant cause of control loop bandwidth variation in previous typical implementations.        The use of a VGA operating at DC to control the transmitted RF power can avoid the levelled output RF power drift due to temperature and voltage variation typically experienced in RF circuits.        
A result of these is that, provided the practical implementation has sufficient control loop bandwidth and dynamic range, the system is functionally equivalent to the idealised circuit shown in FIG. 9 below, with the control reference filter equivalent to the closed-loop transfer function. In bursted TDMA applications, this results in the ability to profile the transmitted RF signal using a rectangular pulse with height proportional to transmitted RF voltage and pulse shaping implemented by a fixed filtering function effected either by the closed-loop transfer function of by an additional dominant filtering function in the control reference connection. In practical mobile handset applications, this means that a single ramp shape can be used together with a scaling multiplication factor to achieve identical ramp shapes at any transmit RF power. This also means there is no need to use different ramp shapes or incorporate any sensitivity switching between frequency bands of operation.
An electronic circuit as proposed is capable of being implemented for power amplifier operation at very high frequencies as an integrated circuit or by other suitable components.
The system can of course be used for non-TDMA applications.
In the circuit shown in FIG. 2, like components to those of FIG. 1 are shown with reference numerals raised by 10. The transmitted signal Ptrans is sampled in a coupler 22, typically realised using a microstrip based directional coupler, and is then multiplied by itself in a mixer circuit or multiplier 28, the multiplying input being derived from the sampled output by an RF limiting amplifier 27. This process results in the production of sum and difference frequency components of which the difference component is a DC voltage equivalent to the full-wave rectification of the sampled RF voltage waveform Ptrans. The DC component is fed to a variable gain amplifier 29 which has a control reference input Vcontrol and an output fed to the power amplifier's control input via a comparator 24 and an integrator 25.
By way of explanation, the output of the mixer circuit 28 is the product of the two inputs:Output=A1·cos(ω·t)·A2·cos(ω·t)
The trigonometric identity for cos (a+b) and cos (a−b) can be used to express the multiplication as a sum and difference of the frequency terms (ωt)
  Output  =                                                        A              1                        ·                          A              2                                2                ⁢                                  ⁢                  cos          ⁡                      (                                          ω                ·                t                            +                              ω                ·                t                                      )                              +                                                  A              1                        ·                          A              2                                2                ⁢                                  ⁢                  cos          ⁡                      (                                          ω                ·                t                            -                              ω                ·                t                                      )                                ⇒                                                      A              1                        ·                          A              2                                2                ⁢                                  ⁢                  cos          ⁡                      (                          2              ·              ω              ·              t                        )                              +                                                  A              1                        ·                          A              2                                2                ⁢                                  ⁢                  cos          ⁡                      (            0            )                              
Thus the multiplier output has a twice input frequency component and a DC component (cos (0)=1). The twice frequency term can be removed with a suitable filter (not shown).
FIG. 3, which illustrates the idealised mixer circuit or multiplier 28, shows that one of the inputs is an output from the limiting amplifier 27 and therefore has an amplitude that is independent of PA output power. Thus the DC component can be represented simply as
  Output  =                    kA        1            2        ⁢                  ⁢          cos      ⁡              (        0        )              ⁢                  ⁢    where    ⁢                  ⁢    k    ⁢                  ⁢    is    ⁢                  ⁢    a    ⁢                  ⁢          constant      .      
In this way the multiplier output can be arranged to be directly proportional to the output power Ptrans of the PA, ie. the power detection circuit has a linear transfer function.